Mixers are needed for frequency conversion in transmitters and receivers and, therefore, belong to the important components for wireless transmission systems. An ideal mixer can be implemented by means of a multiplier. This multiplies a local-oscillator signal by an input signal to be converted to form an output signal.
Conventional mixers are operated within a relatively narrow carrier-frequency range—such as, for example, in WLAN (wireless local area network) applications around 20 MHz. Recently, however, particularly wideband applications are needed in frequency bands from 3 to 10 GHz, so-called UWB (ultra wide band) applications.
FIG. 1 shows a basic modulator or mixer topology.
Two single-sideband mixing branches are provided, the signals OUTU, OUTL of which are combined by means of an adder A1 to form the modulated output signal MOUT. In the branch for the upper sideband, digital input signal data DLF1, which are converted into the analog signal LF1 by a digital/analog converter DA1, are generated from a ROM memory SINROM which contains sinusoidal data, in dependence on a control signal S1. The analog signal LF1 is filtered by a low-pass filter LPF1 to form the signal LH1′ to be converted.
The analog signal LH1′ to be converted is mixed with a sinusoidal carrier sequence signal with the angular frequency ωLO in a multiplicative mixing stage M1 to form the output signal OUTU for the upper sideband.
Analogously, digital data DLF2, an analog conversion signal LF2 for the lower sideband and, by means of a filtered conversion signal LF2′ and a cosinusoidal carrier signal for the lower sideband with the angular frequency ωLO, an output signal OUTL are generated in the branch for the lower sideband by means of a ROM memory COSROM containing cosinusoidal data, a second digital/analog converter DA2, a low-pass filter LPF2 and a multiplicative mixing stage M2 by means of a control signal S2.
A possible multiplicative mixing stage is represented by the so-called Gilbert cell. Such a multiplicative mixing stage according to the prior art is described, for example, in U. Tietze, C H. Schenk: Halbleiter-schaltungstechnik (Semiconductor circuit technology), Edition 12, Springerverlag Berlin, Heidelberg, New York, ISBN 3-540-42849-6.
FIG. 2 shows a corresponding Gilbert cell for multiplicatively mixing a differential input signal LFP, LFN with a differential carrier frequency signal LOP, LON to form a differential output signal OUTP, OUTN.
Accordingly, a first resistor R1, the controlled path of a first transistor T1, the controlled path of a second transistor T2 and the controlled path of a current source transistor T3 are provided in series between a first supply voltage potential VDD and a second supply voltage potential VSS.
A second resistor R2 and the controlled path of a fourth transistor T4 are provided between the first supply voltage potential VDD and the controlled path of the second transistor T2. A bias potential BIAN is applied to a gate terminal of the current source transistor T3.
The controlled path of a fifth transistor T5 and the controlled path of a sixth transistor T6 are connected between the second resistor R2 and the controlled path of the current source transistor T3. Furthermore, the controlled path of a seventh transistor T7 is connected between the first resistor R1 and the controlled path of the sixth transistor T6.
A first component LFP of the differential input signal is connected to the gate terminal of the sixth transistor T6 and the second component LFN of the input signal is connected to the gate terminal of the second transistor T2.
A first component LOP of the carrier frequency signal is connected to the gate terminal of the first transistor T1 and the gate terminal of the fifth transistor T5. The second component LON of the differential carrier frequency signal is connected to the gate terminal of the fourth transistor T4 and the gate terminal of the seventh transistor T7.
The first component OUTP of the differential output signal is picked up between the first resistor and the controlled path of the first transistor T1. The second component OUTN of the output signal is picked up between the second resistor and the controlled path of the fourth transistor T4.
The resistors R1 and R2 act as load resistors. Such a Gilbert cell according to FIG. 2 only has inadequate characteristics of linearity for UWB applications. This leads to harmonic distortions, in particular because the output characteristic of the differential pair of transistors T1, T5, T7, T4 is quadratic.
When they are constructed as NMOS transistors, transistors T1-T7 operate in the inversion region so that the voltages dropped across the load resistors R1, R2 are proportional to the square root of the input voltage which is coupled in by the differential signal LFP, LFN.
With ideal components, this nonlinearity can be largely compensated for by a quadratic predistortion of the input signal LFP, LFN. For this purpose, NMOS diodes can be used, for example, which are connected between the first supply voltage VDD and the respective gate terminals of the transistors T2 and T6. As a result, a quadratic predistorted input signal component in each case occurs which can compensate for the square root nonlinearity of the mixing stage.
In this arrangement, however, accurate matching of the two diodes is very difficult, particularly when the mixing stage is constructed in MOS. A mismatch of such predistortion diodes easily leads to additional components of the carrier frequency in the output signal of such a mixing stage.